Nowadays, many microcircuit LED current stabilizers have appeared, but all of them, as a rule, are quite expensive. And since the need for such stabilizers due to the proliferation of high-power LEDs is great, we have to look for options for making them cheaper.
Here we offer another version of the stabilizer based on the common and cheap MC34063 key stabilizer chip. The proposed version differs from the already known stabilizer circuits on this microcircuit by its slightly non-standard inclusion, which makes it possible to increase the operating frequency and ensure stability even at low values of the inductor inductance and output capacitor capacitance.
Features of the microcircuit - PWM or PWM?
The peculiarity of the microcircuit is that it is both PWM and relay! Moreover, you can choose for yourself what it will be.
The document AN920-D, which describes this microcircuit in more detail, says approximately the following (see the functional diagram of the microcircuit in Fig. 2).
While charging the timing capacitor, a logical one is set at one input of the “AND” logic element that controls the trigger. If the output voltage of the stabilizer is lower than the nominal one (at the input with a threshold voltage of 1.25V), then a logical one is also set at the second input of the same element. In this case, a logical unit is also set at the output of the element and at the input “S” of the trigger, it is set (the active level at the input “S” is logical 1) and at its output “Q” a logical one appears, opening the key transistors.
When the voltage on the frequency-setting capacitor reaches the upper threshold, it begins to discharge, and a logical zero appears at the first input of the “AND” logic element. The same level is supplied to the reset input of the trigger (the active level at the “R” input is logic 0) and resets it. A logical zero appears at the output “Q” of the trigger and the key transistors close.
Then the cycle repeats.
The functional diagram shows that this description applies only to the current comparator, which is functionally connected to the master oscillator (controlled by input 7 of the microcircuit). But the output of the voltage comparator (controlled by input 5) does not have such “privileges”.
It turns out that in each cycle the current comparator can both open the key transistors and close them, if, of course, the voltage comparator allows it. But the voltage comparator itself can only issue permission or prohibition on opening, which can only be processed in the next cycle.
It follows that if you short-circuit the input of the current comparator (pins 6 and 7) and control only the voltage comparator (pin 5), then the key transistors are opened by it and remain open until the end of the capacitor charging cycle, even if the voltage at the comparator input exceeds the threshold. And only when the capacitor begins to discharge will the generator close the transistors. In this mode, the power supplied to the load can only be dosed by the frequency of the master oscillator, since the key transistors, although closed forcibly, are only for a time of the order of 0.3-0.5 μs at any frequency value. And this mode is more similar to PFM - pulse frequency modulation, which belongs to the relay type of regulation.
If, on the contrary, you short-circuit the input of the voltage comparator to the housing, eliminating it from operation, and control only the input of the current comparator (pin 7), then the key transistors will be opened by the master oscillator and closed at the command of the current comparator in each cycle! That is, in the absence of load, when the current comparator does not work, the transistors open for a long time and close for a short period of time. When overloaded, on the contrary, they open and immediately close for a long time at the command of the current comparator. At some average load current values, the keys are opened by the generator, and after some time, after the current comparator is triggered, they are closed. Thus, in this mode, the power in the load is regulated by the duration of the open state of the transistors - that is, full PWM.
It can be argued that this is not PWM, since in this mode the frequency does not remain constant, but changes - it decreases with increasing operating voltage. But with a constant supply voltage, the frequency remains unchanged, and the load current is stabilized only by changing the pulse duration. Therefore, we can assume that this is a full-fledged PWM. And the change in operating frequency when the supply voltage changes is explained by the direct connection of the current comparator with the master oscillator.
When both comparators are used simultaneously (in the classical circuit), everything works exactly the same, and the key mode or PWM is switched on depending on which comparator is triggered at the moment: when there is an overvoltage - the key one (PWM), and when there is an overload on the current - PWM
You can completely eliminate the voltage comparator from operation by shorting the 5th pin of the microcircuit to the housing, and also stabilize the voltage using PWM by installing an additional transistor. This option is shown in Fig. 1.
Fig.1
Voltage stabilization in this circuit is carried out by changing the voltage at the input of the current comparator. The reference voltage is the gate threshold voltage of field-effect transistor VT1. The output voltage of the stabilizer is proportional to the product of the threshold voltage of the transistor and the division coefficient of the resistive divider Rd1, Rd2 and is calculated by the formula:
Uout=Up(1+Rd2/Rd1), where
Up – Threshold voltage VT1 (1.7…2V).
Current stabilization still depends on the resistance of resistor R2.
The operating principle of the current stabilizer.
The MC34063 chip has two inputs that can be used to stabilize the current.
One input has a threshold voltage of 1.25V (5th pin ms), which is not beneficial for fairly powerful LEDs due to power losses. For example, at a current of 700mA (for a 3W LED), we have losses on the current sensor resistor of 1.25*0.7A=0.875W. For this reason alone, the theoretical efficiency of the converter cannot be higher than 3W/(3W+0.875W)=77%. The real one is 60%...70%, which is comparable to linear stabilizers or simply current limiting resistors.
The second input of the microcircuit has a threshold voltage of 0.3V (7th pin ms), and is designed to protect the built-in transistor from overcurrent.
Typically, this is how this microcircuit is used: an input with a threshold of 1.25V - to stabilize voltage or current, and an input with a threshold of 0.3V - to protect the microcircuit from overload.
Sometimes an additional op-amp is installed to amplify the voltage from the current sensor, but we will not consider this option due to the loss of the attractive simplicity of the circuit and the increase in the cost of the stabilizer. It will be easier to take another microcircuit...
In this option, it is proposed to use an input with a threshold voltage of 0.3V to stabilize the current, and simply turn off the other one, with a voltage of 1.25V.
The scheme turns out to be very simple. For ease of perception, the functional units of the microcircuit itself are shown (Fig. 2). 
Fig.2
Purpose and selection of circuit elements.
Diode D with choke L— the elements of any pulse stabilizer are calculated for the required load current and the continuous mode of the inductor current, respectively.
Capacitors Ci and Co– blocking at the entrance and exit. The output capacitor Co is not fundamentally necessary due to small ripples of the load current, especially at large values of the inductor inductance; therefore, it is drawn as a dotted line and may not be present in the real circuit.
Capacitor CT– frequency-setting. It is also not a fundamentally necessary element, so it is shown with a dotted line.
The datasheets for the microcircuit indicate a maximum operating frequency of 100 KHz, the table parameters show an average value of 33 KHz, and the graphs showing the dependence of the duration of the open and closed states of the switch on the capacitance of the frequency-setting capacitor show the minimum values of 2 μs and 0.3 μs, respectively (with a capacitance of 10 pF).
It turns out that if we take the last values, then the period is 2μs+0.3μs=2.3μs, and this is a frequency of 435KHz.
If we take into account the operating principle of the microcircuit - a trigger set by a master oscillator pulse and reset by a current comparator, it turns out that this ms is logical, and the logic has an operating frequency of at least several MHz. It turns out that the performance will be limited only by the speed characteristics of the key transistor. And if it did not operate at a frequency of 400 KHz, then the fronts with pulse decays would be delayed and the efficiency would be very low due to dynamic losses. However, practice has shown that microcircuits from different manufacturers start up well and operate without a frequency-setting capacitor at all. And this made it possible to increase the operating frequency as much as possible - up to 200 KHz - 400 KHz, depending on the type of microcircuit and its manufacturer. The key transistors of the microcircuit maintain such frequencies well, since the pulse rises do not exceed 0.1 μs, and the fall times do not exceed 0.12 μs at an operating frequency of 380 KHz. Therefore, even at such elevated frequencies, dynamic losses in transistors are quite small, and the main losses and heating are determined by the increased saturation voltage of the key transistor (0.5...1V).
Resistor Rb limits the base current of the built-in key transistor. The inclusion of this resistor shown in the diagram allows you to reduce the power dissipated on it and increase the efficiency of the stabilizer. The voltage drop across the resistor Rb is equal to the difference between the supply voltage, the load voltage and the voltage drop across the microcircuit (0.9-2V).
For example, with a series chain of 3 LEDs with a total voltage drop of 9...10V and powered by a battery (12-14V), the voltage drop across the resistor Rb does not exceed 4V.
As a result, the losses on the resistor Rb are several times smaller compared to a typical connection, when the resistor is connected between the 8th pin ms and the supply voltage.
It should be borne in mind that either an additional resistor Rb is already installed inside the microcircuit, or the resistance of the key structure itself is increased, or the key structure is designed as a current source. This follows from the graph of the dependence of the saturation voltage of the structure (between pins 8 and 2) on the supply voltage at various resistances of the limiting resistor Rb (Fig. 3).
Fig.3
As a result, in some cases (when the difference between the supply and load voltages is small or losses can be transferred from resistor Rb to the microcircuit), resistor Rb can be omitted, directly connecting pin 8 of the microcircuit either to the output or to the supply voltage.
And when the overall efficiency of the stabilizer is not particularly important, you can connect pins 8 and 1 of the microcircuit to each other. In this case, the efficiency may decrease by 3-10% depending on the load current.
When choosing the value of the resistor Rb, you have to make a compromise. The lower the resistance, the lower the initial supply voltage the load current stabilization mode begins, but at the same time the losses on this resistor increase over a large range of supply voltage changes. As a result, the efficiency of the stabilizer decreases with increasing supply voltage.
The following graph (Fig. 4), as an example, shows the dependence of the load current on the supply voltage at two different values of the resistor Rb - 24 Ohm and 200 Ohm. It can be clearly seen that with a 200 Ohm resistor, stabilization disappears at supply voltages below 14V (due to insufficient base current of the key transistor). With a 24 Ohm resistor, stabilization disappears at a voltage of 11.5 V.
Fig.4
Therefore, it is necessary to carefully calculate the resistance of the resistor Rb to obtain stabilization in the required range of supply voltages. Especially with battery power, when this range is small and only a few volts.
Resistor Rsc is a load current sensor. The calculation of this resistor has no special features. You should only take into account that the reference voltage of the current input of the microcircuit differs from different manufacturers. The table below shows the actual measured reference voltage values of some microcircuits.
| Chip |
Producer |
U reference (V) |
| MC34063ACD | STMicroelectronics | |
| MC34063EBD | STMicroelectronics | |
| GS34063S | Globaltech Semiconductor | |
| SP34063A | Sipex Corporation | |
| MC34063A | Motorola | |
| AP34063N8 | Analog Technology | |
| AP34063A | Anachip | |
| MC34063A | Fairchild |
Statistics on the value of the reference voltage are small, so the given values should not be considered as a standard. You just need to keep in mind that the actual value of the reference voltage may differ greatly from the value indicated in the datasheet.
Such a large spread in the reference voltage is apparently caused by the purpose of the current input - not load current stabilization, but overload protection. Despite this, the accuracy of maintaining the load current in the above version is quite good.
About sustainability.
The MC34063 chip does not have the ability to introduce correction into the OS circuit. Initially, stability is achieved by increased values of the inductor inductance L and, especially, the capacitance of the output capacitor Co. In this case, a certain paradox arises - when working at higher frequencies, the required pulsations of voltage and load current can be obtained with small inductance and capacitance of the filter elements, but at the same time the circuit can be excited, so it is necessary to install a large inductance and (or) a large capacitance. As a result, the dimensions of the stabilizer are overestimated.
An additional paradox is that for step-down switching stabilizers, the output capacitor is not a fundamentally necessary element. The required level of current (voltage) ripple can be obtained with one choke.
You can obtain good stability of the stabilizer at the required or reduced values of inductance and, especially, output filter capacitance by installing an additional RC correction circuit Rf and Cf, as shown in Figure 2.
Practice has shown that the optimal value of the time constant of this chain should be no less than 1KOhm*uF. Values of chain parameters such as a 10KΩ resistor and a 0.1μF capacitor can be considered quite convenient.
With such a correction circuit, the stabilizer operates stably over the entire supply voltage range, with low values of inductance (units of μH) and capacitance (units and fractions of μF) of the output filter or without an output capacitor at all.
The PWM mode plays an important role in stability when used to stabilize the current input of the microcircuit.
The correction allowed some microcircuits that previously did not want to work normally at all to operate at higher frequencies.
For example, the following graph shows the dependence of the operating frequency on the supply voltage for the MC34063ACD microcircuit from STMicroelectronics with a frequency-setting capacitor capacity of 100 pF.
Fig.5
As can be seen from the graph, without correction this microcircuit did not want to operate at higher frequencies even with a small capacity of the frequency-setting capacitor. Changing the capacitance from zero to several hundred pF did not fundamentally affect the frequency, and its maximum value barely reaches 100 KHz.
After the introduction of the RfCf correction chain, this same microcircuit (like others similar to it) began to operate at frequencies up to almost 300 KHz.
The above dependence can perhaps be considered typical for most microcircuits, although microcircuits from some companies operate at higher frequencies without correction, and the introduction of correction made it possible to obtain for them an operating frequency of 400 KHz at a supply voltage of 12...14V.
The following graph shows the operation of the stabilizer without correction (Fig. 6).
Fig.6
The graph shows the dependences of the consumed current (Ip), load current (In) and output short-circuit current (Isc) on the supply voltage for two values of output capacitor capacitance (Co) - 10 µF and 220 µF.
It is clearly seen that increasing the capacitance of the output capacitor increases the stability of the stabilizer - the broken curves at a capacitance of 10 μF are caused by self-excitation. At supply voltages up to 16V there is no excitation; it appears at 16-18V. Then some kind of mode change occurs and at a voltage of 24V a second kink appears. At the same time, the operating frequency changes, which is also visible in the previous graph (Fig. 5) of the dependence of the operating frequency on the supply voltage (both graphs were obtained simultaneously when examining one instance of the stabilizer).
Increasing the output capacitor capacity to 220 µF or more increases stability, especially at low supply voltages. But it doesn't eliminate the excitement. More or less stable operation of the stabilizer can be achieved with an output capacitor capacity of at least 1000 µF.
In this case, the inductance of the inductor has very little effect on the overall picture, although it is obvious that increasing the inductance increases stability.
Changes in operating frequency affect the stability of the load current, which is also visible in the graph. The overall stability of the output current when the supply voltage changes is also not satisfactory. The current can be considered relatively stable in a fairly narrow range of supply voltages. For example, when running on battery power.
The introduction of the RfCf correction chain radically changes the operation of the stabilizer.
The following graph shows the operation of the same stabilizer but with the RfCf correction chain.
Fig.7
It is clearly visible that the stabilizer began to work as it should be for a current stabilizer - the load and short circuit currents are almost equal and constant over the entire range of supply voltages. In this case, the output capacitor generally ceased to influence the operation of the stabilizer. Now the capacitance of the output capacitor only affects the level of ripple current and voltage of the load, and in many cases the capacitor can not be installed at all.
Below, as an example, are given the values of load current ripple at different capacities of the output capacitor Co. LEDs are connected 3 in series in 10 parallel groups (30 pcs.). Supply voltage - 12V. Choke 47 µH.
Without capacitor: load current 226mA +-65mA or 22.6mA +-6.5mA per LED.
With a 0.33uF capacitor: 226mA +-25mA or 22.6mA +-2.5mA per LED.
With a 1.5uF capacitor: 226mA +-5mA or 22.6mA +-0.5mA per LED.
With a 10uF capacitor: 226mA +-2.5mA or 22.6mA +-0.25mA per LED.
That is, without a capacitor, with a total load current of 226 mA, the load current ripple was 65 mA, which, in terms of one LED, gives an average current of 22.6 mA and a ripple of 6.5 mA.
It can be seen how even a small capacitance of 0.33 μF sharply reduces current ripple. At the same time, increasing the capacitance from 1 µF to 10 µF already has little effect on the ripple level.
All capacitors were ceramic, since conventional electrolytes or tantalum do not provide even close ripple levels.
It turns out that a 1 µF capacitor at the output is quite sufficient for all occasions. Increasing the capacitance to 10 µF with a load current of 0.2-0.3 A hardly makes sense, since the ripple no longer decreases significantly compared to 1 µF.
If you take the inductor with a higher inductance, then you can do without a capacitor even at high load currents and (or) high supply voltages.
The ripple of the input voltage with a 12V supply and the capacity of the input capacitor Ci 10 μF does not exceed 100 mV.
Power capabilities of the microcircuit.
The MC34063 microcircuit operates normally at a supply voltage from 3V to 40V according to datasheets (MS from STM - up to 50V) and up to 45V in reality, providing a load current of up to 1A for a DIP-8 package and up to 0.75A for an SO-8 package. By combining serial and parallel connection of LEDs, you can build a lamp with an output power from 3V*20mA=60mW to 40V*0.75...1A=30...40W.
Taking into account the saturation voltage of the key transistor (0.5...0.8V) and the permissible power of 1.2W dissipated by the microcircuit case, the load current can be increased up to 1.2W/0.8V=1.5A for a DIP-8 package and up to 1A for an SO-8 package.
However, in this case, a good heat sink is required, otherwise the overheating protection built into the chip will not allow operation at such a current.
Standard DIP soldering of the microcircuit body into the board does not provide the required cooling at maximum currents. It is necessary to mold the DIP housing pins for the SMD version, removing the thin ends of the pins. The remaining wide part of the pins is bent flush with the base of the case and only then soldered onto the board. It is useful to position the printed circuit board so that there is a wide area under the microcircuit body, and before installing the microcircuit you need to apply a little thermal conductive paste to its base.
Due to the short and wide pins, as well as due to the tight fit of the housing to the copper polygon of the printed circuit board, the thermal resistance of the microcircuit body is reduced and it will be able to dissipate slightly more power.
For the SO-8 case, installing an additional radiator in the form of a plate or other profile directly on the top of the case helps.
On the one hand, such attempts to increase power look strange. After all, you can simply switch to another, more powerful microcircuit or install an external transistor. And at load currents of more than 1.5A this will be the only the right decision. However, when a load current of 1.3A is required, you can simply improve the heat dissipation and try using a cheaper and simpler option on the MC34063 chip.
The maximum efficiency obtained in this version of the stabilizer does not exceed 90%. Further increase in efficiency is prevented by the increased saturation voltage of the key transistor - at least 0.4...0.5V at currents up to 0.5A and 0.8...1V at currents 1...1.5A. Therefore, the main heating element of the stabilizer is always the microcircuit. True, noticeable heating occurs only at the maximum power for a particular case. For example, a microcircuit in an SO-8 package heats up to 100 degrees at a load current of 1A and, without an additional heat sink, is cyclically turned off by the built-in overheating protection. At currents up to 0.5A...0.7A the microcircuit is slightly warm, and at currents 0.3...0.4A it does not heat up at all.
At higher load currents, the operating frequency can be reduced. In this case, the dynamic losses of the key transistor are significantly reduced. The overall power loss and case heating are reduced.
External elements that affect the efficiency of the stabilizer are diode D, inductor L and resistors Rsc and Rb. Therefore, the diode should be selected with a low forward voltage (Schottky diode), and the inductor should be selected with the winding resistance as low as possible.
You can reduce the losses on the resistor Rsc by reducing the threshold voltage by choosing a microcircuit from the appropriate manufacturer. This has already been discussed earlier (see the table at the beginning).
Another option for reducing losses on the resistor Rsc is to introduce an additional constant current bias of the resistor Rf (this will be shown in more detail below in specific example stabilizer).
Resistor Rb should be carefully calculated, trying to take it with as much resistance as possible. When the supply voltage changes within large limits, it is better to replace the resistor Rb with a current source. In this case, the increase in losses with increasing supply voltage will not be so sharp.
When all of the above measures are taken, the share of losses of these elements is 1.5-2 times less than the losses on the microcircuit.
Since a constant voltage is supplied to the current input of the microcircuit, proportional only to the load current, and not, as usual, a pulse voltage proportional to the current of the key transistor (the sum of the load currents and the output capacitor), the inductance of the inductor no longer affects the stability of operation, since it ceases to be an element correction chain (its role is played by the RfCf chain). Only the amplitude of the key transistor current and the ripple of the load current depend on the inductance value. And since the operating frequencies are relatively high, even with low inductance values the load current ripple is small.
However, due to the relatively low-power key transistor built into the microcircuit, the inductor inductance should not be greatly reduced, since this increases the peak current of the transistor while its average value remains the same and the saturation voltage increases. As a result, the losses on the transistor increase and the overall efficiency decreases.
True, not dramatically - by a few percent. For example, replacing the inductor from 12 µH to 100 µH made it possible to increase the efficiency of one of the stabilizers from 86% to 90%.
On the other hand, this allows, even at low load currents, to choose a choke with low inductance, making sure that the current amplitude of the key transistor does not exceed the maximum value allowed for the microcircuit, 1.5A.
For example, with a load current of 0.2A with a voltage of 9...10V, a supply voltage of 12...15V and an operating frequency of 300KHz, a choke with an inductance of 53µH is required. In this case, the pulse current of the key transistor of the microcircuit does not exceed 0.3A. If we reduce the inductance of the inductor to 4 μH, then at the same average current, the pulse current of the key transistor will increase to the limit value (1.5A). True, the efficiency of the stabilizer will decrease due to increased dynamic losses. But perhaps in some cases it will be acceptable to sacrifice efficiency, but use a small-sized inductor with small inductance.
Increasing the inductance of the inductor also allows you to increase maximum current load up to the maximum current value of the key transistor of the microcircuit (1.5A).
As the inductor inductance increases, the current shape of the switching transistor changes from completely triangular to completely rectangular. And since the area of the rectangle is 2 times larger than the area of the triangle (with the same height and base), the average value of the transistor current (and load) can be increased by 2 times with a constant amplitude of the current pulses.
That is, with a triangular pulse shape with an amplitude of 1.5A, the average current of the transistor and load is:
where k is the maximum pulse duty cycle equal to 0.9 for a given microcircuit.
As a result, the maximum load current does not exceed:
In=1.5A/2*0.9=0.675A.
And any increase in load current above this value entails exceeding the maximum current of the key transistor of the microcircuit.
Therefore, all datasheets for this microcircuit indicate a maximum load current of 0.75A.
By increasing the inductance of the inductor so that the transistor current becomes rectangular, we can remove the two from the maximum current formula and get:
In=1.5A*k=1.5A*0.9=1.35A.
It should be taken into account that with a significant increase in the inductance of the inductor, its dimensions also increase slightly. However, sometimes it turns out to be easier and cheaper to increase the load current by increasing the size of the inductor than installing an additional powerful transistor.
Naturally, with the required load currents of more than 1.5A, there is no way around installing an additional transistor (or another controller microcircuit), and if you are faced with a choice: a load current of 1.4A or another microcircuit, then you should first try to solve the problem by increasing the inductance by going to increasing the throttle size.
The datasheets for the chip indicate that the maximum duty cycle does not exceed 6/7 = 0.857. In reality, values of almost 0.9 are obtained even at high operating frequencies of 300-400 KHz. At lower frequencies (100-200KHz) the duty cycle can reach 0.95.
Therefore, the stabilizer works normally with a small input-output voltage difference.
The stabilizer works interestingly when the load currents are lower than the rated ones, caused by a decrease in the supply voltage below the specified one - the efficiency is at least 95%...
Since PWM is implemented not in the classical way (full control of the master oscillator), but in a “relay” way, using a trigger (start by the generator, reset by the comparator), then at a current below the rated one, a situation is possible when the key transistor stops closing. The difference between the supply and load voltages is reduced to the saturation voltage of the switching transistor, which usually does not exceed 1V at currents up to 1A and no more than 0.2-0.3V at currents up to 0.2-0.3A. Despite the presence of static losses, there are no dynamic ones and the transistor works almost like a jumper.
Even when the transistor remains controlled and operates in PWM mode, the efficiency remains high due to the reduction in current. For example, with a difference of 1.5V between the supply voltage (10V) and the voltage across the LEDs (8.5V), the circuit continued to operate (though at a frequency reduced by half) with an efficiency of 95%.
The current and voltage parameters for this case will be indicated below when considering practical stabilizer circuits.
Practical stabilizer options.
There won’t be many options, since the simplest, repeating classic options according to the circuit design, they do not allow either raising the operating frequency or current, or increasing the efficiency, or obtaining good stability. For this reason the most best option the result is one, the block diagram of which was shown in Fig. 2. Only the component ratings can change depending on the required characteristics of the stabilizer.
Figure 8 shows a diagram of the classic version.
Fig.8
One of the features is that after removing the current of the output capacitor (C3) from the OS circuit, it became possible to reduce the inductance of the inductor. For the test, an old domestic choke on a DM-3 rod with 12 μH was taken. As you can see, the characteristics of the circuit turned out to be quite good.
The desire to increase efficiency led to the circuit shown in Fig. 9
Fig.9
Unlike the previous circuit, resistor R1 is connected not to the power source, but to the output of the stabilizer. As a result, the voltage across resistor R1 became less by the amount of voltage across the load. With the same current through it, the power released on it decreased from 0.5 W to 0.15 W.
At the same time, the inductance of the inductor was increased, which also increases the efficiency of the stabilizer. As a result, efficiency increased by several percent. Specific numbers are shown in the diagram.
Another characteristic feature of the last two schemes. The circuit in Fig. 8 has very good stability of the load current when the supply voltage changes, but the efficiency is rather low. The circuit in Fig. 9, on the contrary, has a fairly high efficiency, but the current stability is poor - when the supply voltage changes from 12V to 15V, the load current increases from 0.27A to 0.3A.
This is not caused the right choice resistance of resistor R1, as mentioned earlier (see Fig. 4). Since the increased resistance R1, reducing the stability of the load current, increases the efficiency, in some cases this can be used. For example, with battery power, when the limits of voltage change are small, and high efficiency is more relevant.
A certain pattern should be noted.
Quite a lot of stabilizers were manufactured (almost all of them were used to replace incandescent lamps with LED lamps in the car interior), and while stabilizers were required from time to time, microcircuits were taken from faulty boards of network “Hubs” and “Switches”. Despite the difference in manufacturers, almost all microcircuits made it possible to obtain decent stabilizer characteristics even in simple circuits.
The only chip I came across was the GS34063S from Globaltech Semiconductor, which in no way wanted to operate at high frequencies.
Then several microcircuits MC34063ACD and MC34063EBD from STMicroelectronics were purchased, which showed even worse results - they did not work at higher frequencies, poor stability, high voltage of the current comparator support (0.45-0.5V), poor stabilization of the load current with good efficiency or poor efficiency with good stabilization...
Perhaps the poor performance of the listed microcircuits is explained by their cheapness - the cheapest ones that were available were purchased, since the MC34063A (DIP-8) microcircuit from the same company, removed from a faulty Switch, worked normally. True, at a relatively low frequency - no more than 160 KHz.
The following microcircuits, taken from broken equipment, worked well:
Sipex Corporation (SP34063A),
Motorola (MC34063A),
Analog Technology (AP34063N8),
Anachip (AP34063 and AP34063A).
Fairchild (MC34063A) - I'm not sure I identified the company correctly.
ON Semiconductor, Unisonic Technologies (UTC) and Texas Instruments - I don’t remember, since I began to pay attention to the company only after I was faced with the reluctance of some companies to work with MS, and I did not specifically buy microcircuits from these companies.
In order not to throw away the purchased, poorly performing MC34063ACD and MC34063EBD microcircuits from STMicroelectronics, several experiments were carried out, which led to the circuit shown at the very beginning in Fig. 2.
The following Fig. 10 shows a practical circuit of a stabilizer with a correction circuit RfCf (in this circuit R3C2). The difference in the operation of the stabilizer without and with a correction chain was already described earlier in the section “On stability” and graphs were presented (Fig. 5, Fig. 6, Fig. 7).
Fig.10
From the graph in Fig. 7 it can be seen that current stabilization is excellent over the entire range of supply voltages of the microcircuit. The stability is very good - as if PWM is working. The frequency is quite high, which makes it possible to use small-sized chokes with low inductance and completely eliminate the output capacitor. Although installing a small capacitor can completely eliminate load current ripple. The dependence of the load current ripple amplitude on the capacitor capacity was discussed earlier in the section “On stability”.
As already mentioned, the MC34063ACD and MC34063EBD microcircuits from STMicroelectronics that I received turned out to have an overestimated reference voltage of the current comparator - 0.45V-0.5V, respectively, despite the value indicated in the datasheet of 0.25V-0.35V. Because of this, at high load currents, large losses occur on the current sensor resistor. To reduce losses, a current source was added to the circuit using transistor VT1 and resistor R2. (Fig. 11).
Fig.11
Thanks to this current source, an additional bias current of 33 μA flows through resistor R3, so the voltage across resistor R3, even without load current, is 33 μA * 10 KΩ = 330 mV. Since the threshold voltage of the current input of the microcircuit is 450 mV, then for the current comparator to operate, the current sensor resistor R1 must have a voltage of 450 mV-330 mV = 120 mV. With a load current of 1A, resistor R1 should be at 0.12V/1A=0.12Ohm. We set the available value to 0.1 Ohm.
Without a current stabilizer on VT1, resistor R1 would have to be selected at the rate of 0.45V/1A=0.45Ohm, and the power would be dissipated on it at 0.45W. Now, at the same current, the loss on R1 is only 0.1 W
This option is powered by a battery, load current up to 1A, power 8-10W. Output short circuit current 1.1A. In this case, the current consumption decreases to 64 mA at a supply voltage of 14.85 V, respectively, the power consumption drops to 0.95 W. The microcircuit does not even heat up in this mode and can remain in short circuit mode as long as desired.
The remaining characteristics are shown in the diagram.
The microcircuit is taken in an SO-8 package and the load current for it is 1A. It gets very hot (terminal temperature is 100 degrees!), so it is better to install the microcircuit in a DIP-8 package converted for SMD mounting, make large polygons and (or) come up with a heatsink.
The saturation voltage of the microcircuit key is quite high - almost 1V at a current of 1A, which is why the heating is so high. Although, judging by the datasheet for the microcircuit, the saturation voltage of the key transistor at a current of 1A should not exceed 0.4V.
Service functions.
Despite the absence of any service capabilities in the microcircuit, they can be implemented independently. Typically, an LED current stabilizer requires switching off and adjusting the load current.
On-off
The stabilizer on the MC34063 chip is turned off by applying voltage to the 3rd pin. An example is shown in Fig.12. 
Fig.12
It was experimentally determined that when voltage is applied to the 3rd pin of the microcircuit, its master oscillator stops and the key transistor closes. In this state, the current consumption of the microcircuit depends on its manufacturer and does not exceed the no-load current specified in the datasheet (1.5-4mA).
Other options for turning off the stabilizer (for example, by applying a voltage of more than 1.25V to the 5th pin) turn out to be worse, since they do not stop the master oscillator and the microcircuit consumes more current compared to control at the 3rd pin.
The essence of such management is as follows.
At the 3rd pin of the microcircuit there is a sawtooth voltage of charge and discharge of the frequency-setting capacitor. When the voltage reaches the threshold value of 1.25V, the capacitor discharge begins and the output transistor of the microcircuit closes. This means that to turn off the stabilizer, you need to apply a voltage of at least 1.25V to the 3rd input of the microcircuit.
According to the datasheets for the microcircuit, the timing capacitor is discharged with a maximum current of 0.26 mA. This means that when an external voltage is applied to the 3rd pin through a resistor, to obtain a switching voltage of at least 1.25V, the current through the resistor must be at least 0.26mA. As a result, we have two main figures for calculating the external resistor.
For example, if the stabilizer supply voltage is 12...15V, the stabilizer must be reliably turned off at the minimum value - at 12V.
As a result, the resistance of the additional resistor is found from the expression:
R=(Up-Uvd1-1.25V)/0.26mA=(12V-0.7V-1.25V)/0.26mA=39KOhm.
To reliably turn off the microcircuit, select the resistor resistance less than the calculated value. In the fragment of the circuit Fig. 12, the resistor resistance is 27KOhm. With this resistance, the turn-off voltage is about 9V. This means that if the stabilizer supply voltage is 12V, you can hope to reliably turn off the stabilizer using this circuit.
When controlling the stabilizer from a microcontroller, resistor R must be recalculated for a voltage of 5V.
The input resistance at the 3rd input of the microcircuit is quite large and any connection of external elements can affect the formation of a sawtooth voltage. To decouple the control circuits from the microcircuit and thereby maintain the same noise immunity, diode VD1 is used.
The stabilizer can be controlled either by applying a constant voltage to the left terminal of resistor R (Fig. 12), or by short-circuiting the connection point between resistor R and diode VD1 to the body (with constant voltage present at the left terminal of resistor R).
Zener diode VD2 is designed to protect the input of the microcircuit from high voltage. At low supply voltages it is not needed.
Load current adjustment
Since the reference voltage of the microcircuit current comparator is equal to the sum of the voltages on resistors R1 and R3, by changing the bias current of resistor R3, the load current can be adjusted (Fig. 11).
Two adjustment options are possible - variable resistor and constant voltage.
Figure 13 shows a fragment of the diagram in Figure 11 with the necessary changes and design relationships that allow you to calculate all the elements of the control circuit.
Fig.13
To regulate the load current with a variable resistor, you need to replace the constant resistor R2 with an assembly of resistors R2’. In this case, when the resistance of the variable resistor changes, the total resistance of resistor R2’ will change within 27...37KOhm, and the drain current of transistor VT1 (and resistor R3) will change within 1.3V/27...37KOhm=0.048...0.035mA. In this case, the bias voltage across resistor R3 will vary within 0.048...0.035mA*10KOhm=0.48...0.35V. To trigger the current comparator of the microcircuit, the voltage on the resistor-current sensor R1 (Fig. 11) must drop 0.45-0.48...0.35V=0...0.1V. With resistance R1=0.1Ohm, such a voltage will drop across it when a load current flows through it in the range of 0…0.1V/0.1Ohm=0…1A.
That is, by changing the resistance of the variable resistor R2’ within 27...37KOhm we can regulate the load current within 0...1A.
To regulate the load current with a constant voltage, you need to install a voltage divider Rd1Rd2 in the gate of transistor VT1. Using this divider, you can match any control voltage with the one required for VT1.
Figure 13 shows all the formulas needed for the calculation.
For example, it is required to regulate the load current within 0...1A using a constant voltage variable within 0...5V.
To use the current stabilizer circuit in Fig. 11, we install a voltage divider Rd1Rd2 in the gate circuit of transistor VT1 and calculate the resistor values.
Initially, the circuit is designed for a load current of 1A, which is set by the current of resistor R2 and the threshold voltage of field-effect transistor VT1. To reduce the load current to zero, as follows from the previous example, you need to increase the current of resistor R2 from 0.034 mA to 0.045 mA. With a constant resistance of resistor R2 (39KOhm), the voltage across it should vary within 0.045…0.034mA*39KOhm=1.755…1.3V. When the gate voltage is zero and the threshold voltage of transistor VT2 is 1.3V, a voltage of 1.3V is set on resistor R2. To increase the voltage on R2 to 1.755V, you need to apply a constant voltage of 1.755V-1.3V=0.455V to gate VT1. According to the conditions of the problem, such a voltage at the gate should be at a control voltage of +5V. Having set the resistance of resistor Rd2 to 100KOhm (to minimize the control current), we find the resistance of resistor Rd1 from the ratio Uу=Ug*(1+Rd2/Rd1):
Rd1= Rd2/(Uу/Ug-1)=100KOhm/(5V/0.455V-1)=10KOhm.
That is, when the control voltage changes from zero to +5V, the load current will decrease from 1A to zero.
Full circuit diagram A 1A current stabilizer with on/off and current adjustment functions is shown in Fig. 14. The numbering of new elements continues what was started according to the scheme in Fig. 11.
Fig.14
The circuit was not tested as part of Fig. 14. But the circuit according to Fig. 11, on the basis of which it was created, was fully tested.
The on/off method shown in the diagram has been tested by prototyping. Current control methods have so far been tested only by simulation. But since the adjustment methods are created on the basis of a really proven current stabilizer, during assembly you only have to recalculate the resistor values to match the parameters of the applied field-effect transistor VT1.
In the above circuit, both options for adjusting the load current are used - with a variable resistor Rp and a constant voltage of 0...5V. The adjustment with a variable resistor was chosen slightly differently compared to Fig. 12, which made it possible to apply both options simultaneously.
Both adjustments are dependent - the current set in one way is the maximum for the other. If the variable resistor Rp is used to set the load current to 0.5A, then by adjusting the voltage the current can be changed from zero to 0.5A. And vice versa - a current of 0.5A, set by a constant voltage, with a variable resistor will also change from zero to 0.5A.
The dependence of the load current adjustment by a variable resistor is exponential, therefore, to obtain linear adjustment, it is advisable to select a variable resistor with a logarithmic dependence of the resistance on the angle of rotation.
As resistance Rp increases, the load current also increases.
The dependence of load current regulation by constant voltage is linear.
Switch SB1 turns the stabilizer on or off. When the contacts are open, the stabilizer is turned off, when contacts are closed, it is on.
With fully electronic control, turning off the stabilizer can be achieved either by applying a constant voltage directly to the 3rd pin of the microcircuit, or by means of an additional transistor. Depending on the required control logic.
Capacitor C4 ensures a soft start of the stabilizer. When power is applied, until the capacitor is charged, the current of field-effect transistor VT1 (and resistor R3) is not limited by resistor R2, but is equal to the maximum for the field-effect transistor turned on in current source mode (units - tens of mA). The voltage across resistor R3 exceeds the threshold for the current input of the microcircuit, so the key transistor of the microcircuit is closed. The current through R3 will gradually decrease until it reaches the value set by resistor R2. As this value approaches, the voltage on resistor R3 decreases, the voltage at the current protection input increasingly depends on the voltage on the current sensor resistor R1 and, accordingly, on the load current. As a result, the load current begins to increase from zero to a predetermined value (by a variable resistor or a constant control voltage).
Printed circuit board.
Below are options for the stabilizer printed circuit board (according to the block diagram of Fig. 2 or Fig. 10 - a practical version) for different chip packages (DIP-8 or SO-8) and different chokes (standard, factory-made or homemade on a sprayed iron ring ). The board was drawn in the Sprint-Layout program version 5:
All options are designed for installation of SMD elements of standard sizes from 0603 to 1206, depending on the calculated power of the elements. The board has seats for all elements of the circuit. When desoldering the board, some elements may not be installed (this has already been discussed above). For example, I have already completely abandoned the installation of frequency-setting C T and output Co capacitors (Fig. 2). Without a frequency-setting capacitor, the stabilizer operates at a higher frequency, and the need for an output capacitor is only at high load currents (up to 1A) and (or) small inductances of the inductor. Sometimes it makes sense to install a frequency-setting capacitor, reducing the operating frequency and, accordingly, dynamic power losses at high load currents.
Any features printed circuit boards do not have and can be made on both single-sided and double-sided foil PCB. When using double-sided PCB, the second side is not etched and serves as an additional heat sink and (or) a common wire.
When using metallization on the back side of the board as a heat sink, you need to drill a through hole near the 8th pin of the microcircuit and solder both sides together with a short jumper made of thick copper wire. If you use a microcircuit in a DIP package, then the hole must be drilled against the 8th pin and when soldering, use this pin as a jumper, soldering the pin on both sides of the board.
Instead of a jumper, good results are achieved by installing a rivet made of copper wire with a diameter of 1.8 mm (a cable core with a cross-section of 2.5 mm2). The rivet is placed immediately after etching the board - you need to drill a hole with a diameter equal to the diameter of the rivet wire, insert a piece of wire tightly and shorten it so that it protrudes from the hole no more than 1 mm, and rivet it thoroughly on both sides on the anvil with a small hammer. On the installation side, the rivet should be flush with the board so that the protruding head of the rivet does not interfere with the unsoldering of the parts.
It may seem strange advice to make a heat sink specifically from the 8th pin of the microcircuit, but a crash test of the case of a faulty microcircuit showed that its entire power part is located on a wide copper plate with a solid outlet to the 8th pin of the case. Pins 1 and 2 of the microcircuit, although made in the form of strips, are too thin to be used as a heat sink. All other terminals of the case are connected to the microcircuit crystal with thin wire jumpers. Interestingly, not all microcircuits are designed this way. Several more cases tested showed that the crystal is located in the center, and the strip pins of the microcircuit are all the same. Wiring - with wire jumpers. Therefore, to check it, you need to “disassemble” several more microcircuit housings...
The heat sink can also be made from a copper (steel, aluminum) rectangular plate 0.5-1 mm thick with dimensions that do not extend beyond the board. When using a DIP package, the plate area is limited only by the height of the inductor. You should put a little thermal paste between the plate and the chip body. With an SO-8 package, some mounting parts (capacitors and diode) can sometimes prevent a tight fit of the plate. In this case, instead of thermal paste, it is better to use a Nomakon rubber gasket of suitable thickness. It is advisable to solder the 8th pin of the microcircuit to this plate with a jumper wire.
If the cooling plate is large and blocks direct access to the 8th pin of the microcircuit, then you need to first drill a hole in the plate opposite the 8th pin, and first solder a piece of wire vertically to the pin itself. Then, thread the wire through the hole in the plate and press it against the chip body, solder them together.
A good flux for soldering aluminum is now available, so it is better to make a heat sink from it. In this case, the heat sink can be bent along the profile with the largest surface area.
To obtain load currents of up to 1.5A, the heat sink should be made on both sides - in the form of a solid polygon on the back side of the board and in the form of a metal plate pressed against the chip body. In this case, it is necessary to solder the 8th pin of the microcircuit both to the polygon on the back side and to the plate pressed to the case. To increase the thermal inertia of the heat sink on the back side of the board, it is also better to make it in the form of a plate soldered to the polygon. In this case, it is convenient to place the heat-sinking plate on the rivet at the 8th pin of the microcircuit, which previously connected both sides of the board. Solder the rivet and plate, and secure it with soldering in several places around the perimeter of the board.
By the way, when using a plate on the back side of the board, the board itself can be made of one-sided foil PCB.
The inscriptions on the board for the positional designations of the elements are made in the usual way (as are the printed tracks), except for the inscriptions on the polygons. The latter are made on a white service layer “F”. In this case, these inscriptions are obtained by etching.
The power and LED wires are soldered at opposite ends of the board according to the inscriptions: “+” and “-” for power, “A” and “K” for LEDs.
When using the board in an uncased version (after checking and tuning), it is convenient to thread it into a piece of heat-shrink tubing of a suitable length and diameter and heat it with a hairdryer. The ends of the heat shrink that has not yet cooled down must be crimped with pliers closer to the terminals. The hot-pressed heat shrink glues together and forms an almost airtight and fairly durable housing. The crimped edges are glued so tightly that when you try to separate, the heat shrink simply breaks. At the same time, if repair or maintenance is necessary, the crimped areas unstick themselves when reheated with a hairdryer, without leaving even traces of crimping. With some skill, you can stretch the still hot heat shrink with tweezers and carefully remove the board from it. As a result, the heat shrink will be suitable for re-packaging the board.
If it is necessary to completely seal the board, after compressing the thermal pad, its ends can be filled with thermal pad. To strengthen the “case”, you can put two layers of heat shrink on the board. Although one layer is quite durable.
Stabilizer calculation program
To quickly calculate and evaluate the elements of the circuit, a table with formulas was drawn in the EXCEL program. For convenience, some calculations are supported by VBA code. The operation of the program was tested only in Windows XP:
When you run the file, a window may appear warning you about the presence of macros in the program. You should select the “Don’t disable macros” command. Otherwise, the program will start and even perform recalculation using the formulas written in the table cells, but some functions will be disabled (checking the correctness of the input, the ability to optimize, etc.).
After starting the program, a window will appear asking: “Restore all input data to default?” In which you need to click the “Yes” or “No” button. If you select “Yes”, all input data for the calculation will be set by default, as an example. All calculation formulas will also be updated. If you select "No", the input data will use the values saved in the previous session.
Basically, you need to select the “No” button, but if you do not want to save the previous calculation results, you can select “Yes”. Sometimes, if you enter too many incorrect input data, some kind of malfunction, or accidentally delete the contents of a cell with a formula, it is easier to exit the program and run it again by answering the question “Yes”. This is easier than searching for and correcting errors and re-prescribing lost formulas.
The program is a regular Excel worksheet with three separate tables ( Input data , Output , Calculation results ) and stabilizer circuit.
The first two tables contain the name of the entered or calculated parameter, its short symbol (it is also used in formulas for clarity), the value of the parameter and the unit of measurement. In the third table, the names are omitted as unnecessary, since the purpose of the element can be seen right there in the diagram. The values of the calculated parameters are marked in yellow and cannot be changed independently, since formulas are written in these cells.
To the table " Input data » the initial data is entered. The purpose of some parameters is explained in the notes. All cells with input data must be filled in, since they all take part in the calculation. The exception is the cell with the parameter “Load current ripple (Inp)” - it may be empty. In this case, the inductance of the inductor is calculated based on the minimum value of the load current. If you set the value of the load ripple current in this cell, then the inductance of the inductor is calculated based on the specified ripple value.
Some parameters may differ among different chip manufacturers - for example, the value of the reference voltage or current consumption. To obtain more reliable calculation results, you need to provide more accurate data. To do this, you can use the second sheet of the file (“Chips”), which contains the main list of different parameters. Knowing the chip manufacturer, you can find more accurate data.
In the table " Output » intermediate calculation results of interest are found. The formulas used for calculations can be seen by selecting the cell with the calculated value. A cell with the “Maximum fill factor (dmax)” parameter can be highlighted in one of two colors – green and red. The cell is highlighted in green when the parameter value is acceptable, and in red when the maximum allowable value is exceeded. In the cell note you can read which input data needs to be changed to correct it.
The AN920-D document, which describes this chip in more detail, states that the maximum duty cycle value of the MC34063 chip cannot exceed 0.857, otherwise the control limits may not coincide with the specified ones. It is this value that is taken as the criterion for the correctness of the parameter obtained in the calculation. True, practice has shown that the real value of the fill factor can be greater than 0.9. Apparently, this discrepancy is explained by “non-standard” inclusion.
The result of the calculations is the values of the passive elements of the circuit, summarized in the third table " Calculation results" . The obtained values can be used when assembling the stabilizer circuit.
Sometimes it is useful to adjust the obtained values to suit yourself, for example, when the obtained value of the resistor resistance, capacitor capacitance or inductor inductance does not coincide with the standard one. It is also interesting to see how changing the values of some elements affects the overall characteristics of the circuit. This feature is implemented in the program.
To the right of the table " Calculation results" There is a square next to each parameter. When you click the left mouse button on the selected square, a “bird” appears in it, marking the parameter that requires selection. In this case, the yellow highlight is removed from the field with the value, which means that you can independently select the value of this parameter. And in the table " Input data" The parameters that change are highlighted in red. That is, a reverse recalculation is performed - the formula is written in a cell of the input data table, and the parameter for calculation is the table value " Calculation results" .
For example, by placing a “bird” opposite the inductance of the inductor in the table “ Calculation results" , you can see that the “Minimum load current” parameter of the table “ is highlighted in red Input data ».
When the inductance changes, some parameters of the table also change " Output ", for example, "Maximum inductor and switch current (I_Lmax)". In this way, you can select a choke with the minimum inductance from the standard range and dimensions, without exceeding the maximum current of the key transistor of the microcircuit, but “sacrificing” the value of the minimum load current. At the same time, you can see that the value of the output capacitor Co also increased to compensate for the increase in load current ripple.
Having selected the inductance and made sure that the other dependent parameters do not go beyond dangerous limits, remove the check mark next to the inductance parameter, thereby securing the result obtained before changing other parameters that affect the inductance of the inductor. Moreover, in the table “ Calculation results" formulas are restored, and in the table " Input data" , on the contrary, are removed.
In the same way, you can select other parameters of the table " Calculation results" . However, you should keep in mind that the parameters of almost all formulas overlap, so if you want to change all the parameters of this table at once, an error window may appear with a message about cross-references.
Download the article in pdf format.
Basic specifications MC34063
- Wide range of input voltages: from 3 V to 40 V;
- High output pulse current: up to 1.5 A;
- Adjustable output voltage;
- Converter frequency up to 100 kHz;
- Internal reference accuracy: 2%;
- Short circuit current limitation;
- Low consumption in sleep mode.
- Reference voltage source 1.25 V;
- Comparator comparing the reference voltage and the input signal from input 5;
- Pulse generator resetting RS trigger;
- Element AND combining signals from the comparator and generator;
- RS trigger eliminating high-frequency switching of output transistors;
- Driver transistor VT2, in the emitter follower circuit, to amplify the current;
- Output transistor VT1 provides current up to 1.5A.
MC34063 boost converter
For example, I used this chip to get 12 V power for the interface module from a laptop USB port (5 V), so the interface module worked when the laptop was running; it did not need its own uninterruptible power supply.It also makes sense to use the IC to power contactors, which need a higher voltage than other parts of the circuit.
Although the MC34063 has been produced for a long time, its ability to operate on 3 V allows it to be used in voltage stabilizers powered by lithium batteries.
Let's look at an example of a boost converter from the documentation. This circuit is designed for an input voltage of 12 V, an output voltage of 28 V at a current of 175 mA.
- C1 – 100 µF 25 V;
- C2 – 1500 pF;
- C3 – 330 µF 50 V;
- DA1 – MC34063A;
- L1 – 180 µH;
- R1 – 0.22 Ohm;
- R2 – 180 Ohm;
- R3 – 2.2 kOhm;
- R4 – 47 kOhm;
- VD1 – 1N5819.
Buck converter on MC34063
Reducing the voltage is much easier - there are a large number of compensating stabilizers that do not require inductors and require fewer external elements, but for a pulse converter there is work when the output voltage is several times less than the input, or the conversion efficiency is simply important.The technical documentation provides an example of a circuit with an input voltage of 25 V and an output voltage of 5 V at a current of 500 mA.
- C1 – 100 µF 50 V;
- C2 – 1500 pF;
- C3 – 470 µF 10 V;
- DA1 – MC34063A;
- L1 – 220 µH;
- R1 – 0.33 Ohm;
- R2 – 1.3 kOhm;
- R3 – 3.9 kOhm;
- VD1 – 1N5819.
MC34063 inverting converter circuit
The third scheme is used less frequently than the first two, but is no less relevant. Accurate voltage measurements or amplification of audio signals often require bipolar power supply, and the MC34063 can help provide negative voltages.The documentation provides a circuit that allows you to convert a voltage of 4.5 .. 6.0 V into a negative voltage of -12 V with a current of 100 mA.
- C1 – 100 µF 10 V;
- C2 – 1500 pF;
- C3 – 1000 µF 16 V;
- DA1 – MC34063A;
- L1 – 88 µH;
- R1 – 0.24 Ohm;
- R2 – 8.2 kOhm;
- R3 – 953 Ohm;
- VD1 – 1N5819.
Analogues of the MC34063 chip
If MC34063 is intended for commercial applications and has an operating temperature range of 0 .. 70°C, then its full analogue MC33063 can operate in a commercial range of -40 .. 85°C.Several manufacturers produce MC34063, other chip manufacturers produce complete analogues: AP34063, KS34063. Even the domestic industry produced a complete analogue K1156EU5, and although it’s a big problem to buy this microcircuit now, you can find many diagrams of calculation methods specifically for the K1156EU5, which are applicable to the MC34063.
If you need to develop a new device and the MC34063 seems to fit perfectly, then you should pay attention to more modern analogues, for example: NCP3063.
Some time ago I already published a review where I showed how to make a PWM stabilizer using KREN5. Then I mentioned one of the most common and probably the cheapest DC-DC converter controllers. Microcircuit MC34063.
Today I will try to complement the previous review.
In general, this microcircuit can be considered outdated, but nevertheless it enjoys well-deserved popularity. Mainly due to the low price. I still use them sometimes in my various crafts.
That’s actually why I decided to buy myself a hundred of these little things. They cost me 4 dollars, now from the same seller they cost 3.7 dollars per hundred, that’s only 3.7 cents apiece.
You can find them cheaper, but I ordered them as a kit with other parts (reviews of a charger for a lithium battery and a current stabilizer for a flashlight). There is also a fourth component, which I ordered there, but more on that another time.
Well, I’ve probably already bored you with the long introduction, so I’ll move on to the review.
Let me warn you right away, there will be a lot of photos.
It all came in bags, wrapped in bubble wrap. Such a bunch :) 
The microcircuits themselves are neatly packed in a bag with a latch, and a piece of paper with the name is pasted onto it. It was written by hand, but I don’t think there will be any problems recognizing the inscription. 
These microcircuits are produced by different manufacturers and are also labeled differently.
MC34063
KA34063
UCC34063
Etc.
As you can see, only the first letters change, the numbers remain unchanged, which is why it is usually called simply 34063.
I got the first ones, MC34063. 
The photo is next to the same mikruha, but from a different manufacturer.
The one under review stands out with clearer markings. 
I don’t know what else can be seen, so I’ll move on to the second part of the review, the educational one.
DC-DC converters are used in many places; now it is probably difficult to find an electronic device that does not have them.
There are three main conversion schemes, all of them are described in 34063, as well as in its application, and in one more.
All the described circuits do not have galvanic isolation. Also, if you look closely at all three circuits, you will notice that they are very similar and differ in the interchange of three components, the inductor, the diode and the power switch.
First, the most common one.
Step-down or step-down PWM converter.
It is used where it is necessary to reduce the voltage, and to do this with maximum efficiency.
The input voltage is always greater than the output voltage, usually at least 2-3 Volts; the greater the difference, the better (within reasonable limits).
In this case, the current at the input is less than at the output.
This circuit design is often used on motherboards, although the converters there are usually multi-phase and with synchronous rectification, but the essence remains the same, Step-Down.
In this circuit, the inductor accumulates energy when the key is open, and after the key is closed, the voltage across the inductor (due to self-induction) charges the output capacitor 
The next scheme is used a little less frequently than the first.
It can often be found in Power-banks, where a battery voltage of 3-4.2 Volts produces a stabilized 5 Volts.
Using such a circuit, you can get more than 5 Volts, but it must be taken into account that the greater the voltage difference, the harder it is for the converter to work.
There is also one not very pleasant feature of this solution: the output cannot be disabled “software”. Those. The battery is always connected to the output via a diode. Also, in the case of a short circuit, the current will be limited only by the internal resistance of the load and battery.
To protect against this, either fuses or an additional power switch are used.
Just like last time, when the power switch is open, energy is first accumulated in the inductor; after the key is closed, the current in the inductor changes its polarity and, summed with the battery voltage, goes to the output through the diode.
The output voltage of such a circuit cannot be lower than the input voltage minus the diode drop.
The current at the input is greater than at the output (sometimes significantly). 
The third scheme is used quite rarely, but it would be wrong not to consider it.
This circuit has an output voltage of opposite polarity than the input.
It's called an inverting converter.
In principle, this circuit can either increase or decrease the voltage relative to the input, but due to the peculiarities of the circuit design, it is often used only for voltages greater than or equal to the input.
The advantage of this circuit design is the ability to turn off the output voltage by closing the power switch. The first scheme can do this as well.
As in previous schemes, energy is accumulated in the inductor, and after closing the power switch it is supplied to the load through a reverse-connected diode. 
When I conceived this review, I didn’t know what would be better to choose as an example.
There were options to make a step-down converter for PoE or a step-up converter to power an LED, but somehow all this was uninteresting and completely boring.
But a few days ago a friend called and asked me to help him solve a problem.
It was necessary to obtain a stabilized output voltage regardless of whether the input was greater or less than the output.
Those. I needed a buck-boost converter.
The topology of these converters is called (Single-ended primary-inductor converter).
A couple more good documents on this topology. , .
The circuit of this type of converter is noticeably more complex and contains an additional capacitor and inductor. 
This is how I decided to do it
For example, I decided to make a converter capable of producing stabilized 12 Volts when the input fluctuates from 9 to 16 Volts. True, the power of the converter is small, since the built-in key of the microcircuit is used, but the solution is quite workable.
If you make the circuit more powerful, install an additional field-effect transistor, chokes for higher current, etc. then such a circuit can help solve the problem of powering a 3.5-inch hard drive in a car.
Also, such converters can help solve the problem of obtaining, which has already become popular, a voltage of 3.3 Volts from one lithium battery in the range of 3-4.2 Volts.
But first, let's turn the conditional diagram into a principle one. 
After that, we’ll turn it into a trace; we won’t sculpt everything on the circuit board. 
Well, next I will skip the steps described in one of my tutorials, where I showed how to make a printed circuit board.
The result was a small board, the dimensions of the board were 28x22.5, the thickness after sealing the parts was 8mm. 
I dug up all sorts of different parts around the house.
I had chokes in one of the reviews.
There are always resistors.
The capacitors were partially present and partially removed from various devices.
The 10 µF ceramic one was removed from an old hard drive (they are also found on monitor boards), the aluminum SMD one was taken from an old CD-ROM. 
I soldered the scarf and it turned out neat. I should have taken a photo on some matchbox, but I forgot. The dimensions of the board are approximately 2.5 times smaller than a matchbox. 
The board is closer, I tried to arrange the board more tightly, there is not a lot of free space.
A 0.25 Ohm resistor is formed into four 1 Ohm resistors in parallel on 2 levels. 
There are a lot of photos, so I put them under a spoiler
I checked in four ranges, but by chance it turned out to be in five, I didn’t resist this, but simply took another photo.
I didn’t have a 13K resistor, I had to solder it to 12, so the output voltage is somewhat underestimated.
But since I made the board simply to test the microcircuit (that is, this board itself no longer has any value for me) and write a review, I didn’t bother.
The load was an incandescent lamp, the load current was about 225mA
Input 9 Volts, output 11.45 
The input is 11 Volts, the output is 11.44. 
The input is 13 volts, the output is still the same 11.44 
The input is 15 Volts, the output is again 11.44. :) 
After that I thought about finishing it, but since the diagram indicated a range of up to 16 Volts, I decided to check at 16.
At the entrance 16.28, at the exit 11.44 
Since I got hold of a digital oscilloscope, I decided to take oscillograms.
I also hid them under the spoiler, since there are quite a lot of them
This is of course a toy, the power of the converter is ridiculous, although useful.
But I picked up a few more for a friend on Aliexpress.
Perhaps it will be useful for someone. 
- 20.09.2014
A trigger is a device with two stable equilibrium states, designed for recording and storing information. A flip-flop is capable of storing 1 bit of data. Symbol The trigger has the form of a rectangle, inside of which the letter T is written. Input signals are connected to the left of the rectangle image. Designations of signal inputs are written in an additional field on the left side of the rectangle. ...
- 21.09.2014
The single-cycle output stage of a tube amplifier contains a minimum of parts and is easy to assemble and adjust. Pentodes in the output stage can only be used in ultra-linear, triode or normal modes. With triode connection, the shielding grid is connected to the anode through a 100...1000 Ohm resistor. In an ultralinear connection, the cascade is covered by the OS along the shielding grid, which reduces ...
- 04.05.2015
The figure shows a diagram of a simple infrared remote control and a receiver whose executive element is a relay. Due to the simplicity of the remote control circuit, the device can only perform two actions: turn on the relay and turn it off by releasing the S1 button, which may be sufficient for certain purposes (garage doors, opening an electromagnetic lock, etc.). Setting up the circuit is very...
- 05.10.2014
The circuit is made using a dual op-amp TL072. A pre-amplifier with coefficient is made on A1.1. amplification by a given ratio R2\R3. R1 is the volume control. Op amp A1.2 has an active three-band bridge tone control. Adjustments are made by variable resistors R7R8R9. Coef. transmission of this node 1. Charged preliminary ULF supply can be from ±4V to ±15V Literature...
Very often the question arises of how to obtain the voltage required for a power supply circuit, having a source with a different voltage from the required one. Such tasks are divided into two: when: you need to reduce or increase the voltage to a given value. This article will consider the first option.
As a rule, you can use a linear stabilizer, but it will have large power losses, because it will convert the difference in voltage into heat. This is where pulse converters come to the rescue. We present to your attention a simple and compact converter based on the MC34063.
This chip is very versatile, it can implement buck, boost and inverting converters with a maximum internal current of up to 1.5A. But this article only discusses the step-down converter, the rest will be discussed later.
The dimensions of the resulting converter are 21x17x11 mm. Such dimensions were obtained due to the use of lead and SMD parts together. The converter contains only 9 parts.
The parts in the circuit are designed for 5V with a current limit of 500mA, with a ripple of 43kHz and 3mV. The input voltage can be from 7 to 40 volts.
The resistor divider on R2 and R3 is responsible for the output voltage; if you replace them with a trimming resistor of about 10 kOhm, then you can set the required output voltage. Resistor R1 is responsible for limiting the current. Capacitor C1 and coil L1 are responsible for the ripple frequency, and capacitor C3 is responsible for the ripple level. The diode can be replaced with 1N5818 or 1N5820. To calculate the circuit parameters there is a special calculator - http://www.nomad.ee/micros/mc34063a/index.shtml, where you just need to set the required parameters, it can also calculate the circuits and parameters of the two types of converters not considered.
2 printed circuit boards were made: on the left - with a voltage divider on a voltage divider made of two resistors of standard size 0805, on the right - with a variable resistor 3329H-682 6.8 kOhm. The MC34063 chip is in a DIP package, underneath it are two chip tantalum capacitors of standard size - D. Capacitor C1 is of standard size 0805, an output diode, a current limiting resistor R1 - half a watt, at low currents, less than 400 mA, you can install a resistor of lower power. Inductance CW68 22uH, 960mA.
Ripple waveforms, R limit = 0.3 Ohm
These oscillograms show ripples: on the left - without a load, on the right - with a load in the form of a cell phone, limiting a 0.3 Ohm resistor, below with the same load, but limiting a 0.2 Ohm resistor.
Ripple waveform, R limit = 0.2 Ohm
The characteristics taken (not all parameters were measured), with an input voltage of 8.2 V.
This adapter was made to recharge a cell phone and power digital circuits while traveling.
The article showed a board with a variable resistor as a voltage divider, I will add the corresponding circuit to it, the difference from the first circuit is only in the divider.